Frequency modulated wave receiver



Feb. 17, 1942. c. N. KIMBALL. ETAL.

FREQUENCY MODULATED WAVE RECEIVER Filed June 14, 1940 Patented F eb. 17,1942 FREQUENCY ivronntarnn WAVE RECEIVER Charles N. Kimball, EastOrange, N. J., and George C. Sziklai, Bloomington, Ind., assignors toRadio Corporation of America, a corporation of Delaware application.time 1i, i940, serial No. 340,550

3 Claims.

Our present invention relatesto reception of frequency modulated (F. M.)waves, and more for, employing degenerative feedback in an F. M.particularly to a novel method of, and means receiver.

F. M. reception, as is well known, is restricted at the present time tofrequencies of the order of l-50 megacycles (ma), a maximum frequencydeviation of the carrier of i100 kilocycles (kc.) being permitted. Thereare many instances When it is desired to receive amplitude modulatedwaves in the standard broadcast band of 550-1'790 kc. Receivers whichare able to receive the amplitude modulated signals in the standardbroadcast range, as well as the F. M. signals in the short Wave range,would normally have to embody two distinct receiving systems. This isparticularly so since in the standard broadcast range, and in the caseof superheterodyne reception, the intermediate frequency (I. F.) bandwidth is approximately l0 kc. For reception of both F. M. and amplitudemodulated signals in the standard broadcast range it would be highlydesirable to utilize the same I. F. networks, while restricting thechange-over switching operations to the networks prior to the rstdetector and to the discriminator network.

Again, even where receiving solely F. M. Waves, it may be desirable toreceive F. M. waves not only of a 100 kc. frequency deviation, but alsowaves having a frequency deviation of the order of -80 kc. In such casesas well it would be highly desirable to utilize common I. F. networksfor both types of frequency deviation, and to restrict the switchingoperation to the networks preceding the rst detector. For example, inthe last case, it would be most desirable to change over from signalfrequency networks having a band width of say 50 kc. to 200 kc., and yetutilize the succeeding I. F. networks which would retain the 50 kc. bandwidth.

Accordingly, it may be stated to be one of the main objects of ourpresent invention to provide a superheterodyne receiver for F. M. wavesutilizing I. F. networks of a fixed band width value, but Whose pre-rstdetector networks are adapted to be of a substantially greater bandwidth, or if desired, of the same band width asthe I. F. networks; therebeing employed an audio degenerative feedback in the receiving systemwhen the pre-first detector networks are of a substantially greater bandwidth.

Another important object of our present in-l vention is to provide an F.M. superheterodyne receiver whose I. F. networks' are of a substan- (Cl.Z50-20) tially smaller band width than the pre-first detector networks,wherein a portion of the audio output voltage of the second detector isutilized to produce frequency modulation oftheflocal oscillator outputwaves, and the latter frequency modulation being heterodyned so that itsenvelope of frequency variation is in phase with that of the received F.M. sig-naL'so that the resultant I. F. frequency deviation is reduced tothe difference of the frequency deviations 0f the F. M. wave and thefrequency-modulated local oscillations.

AStill another object of this-invention is to provide an audiodegenerative feedback in a superlieterodyne receiver of the F. M. type,wherein as a result of sufficient degenerative feedback to the localoscillator the output of the second detector is substantiallyindependent of the receiver input signal amplitude, thereby resulting inAsuppression of noise and other undesired forms of amplitude modulation,and also effecting a reduction in the amplitude distortionproduced bynon-uniformity in the I. F. response and by discriminator non-linearityover the reduced I. F. band.

In a system of the'type described above, wherein audio degenerativefeedback to the local oscillator is kemployed for reducing the frequencydeviation of the applied F. M. wave, there results an appreciable timedelay in the system, which causes a phase difference between thefrequency deviation envelope appearing at the antenna terminals and thatresulting from modulatingA the local oscillatorfrequency. This phasedifference is negligibly small for low audio frequencies, and increaseswith audio frequency until at some audio frequency the phase delaybetween the antenna input frequency deviation envelope and theenvelopeof the local oscillator is degrees at which point neitherdegeneration or regeneration exists. If this variation of phase delaywith audio modulating frequency is permitted to eX- ist, it will resultin a variation of effective I. F. band width with audio frequency. Itistheoretically possible to compensate for vthe phase delay encountered inthe I. F. circuits by inserting corrective networks in the feedback pathwhich result in a phase advance for audio frequencies which nullies thephase delay in the I. F. networks. It is, also, possible to increase theoscillation frequency deviation greatly vat high audiofrequenciesthereby to produce an essential in-phase degenerative frequencydeviationcomponent to beat with the input signal.

Accordingly, it may be stated to be another object of this invention toprovide phase advancing networks in the degenerative feedback path so asto compensate for the phase delay caused by the resonant I. F. networksof the system.

Still other objects of this invention are to improve generally theefiiciency of superheterodyne receivers adapted for F. M. waves, andmore especially to provide receivers of the latter type which areadapted for flexible operation, and are economically manufactured andassembled.

The novel features which we believe to'be characteristic of ourinvention are set forth in particularity in the appended claims; theinvention itself, however, as to both its organization and method ofoperation will best be understood by reference to the followingdescription taken in connection with the drawing in which we haveindicated diagrammatically a circuit organization whereby our inventionmay be carried into effect.

In the drawing:

Fig. 1 shows an F. M. receiving system embodying the invention,

Fig. 2 shows graphically the second detector characteristic.

Referring now to the accompanying drawing, and particularly to Fig. 1,there is shown an F. M. receiver which is of the superheterodyne type.The conventional networks of this receiver are schematicallyrepresented, since those skilled in the art at the present time arefully aware of the manner of constructing such a receiver. The signalcollector I may be a di-pole, a grounded antenna circuit, or even aradio frequency distribution line. It is assumed that the collector Icollects F. M. signals in a range of approximately 42 to 50 mc.,although it is to be clearly understood that the present invention isnot restricted in any way to that signal frequency range. The collectedsignals are fed to a tunable radio frequency amplifier 2 which mayembody one or more tunable stages of radio frequency amplification. Thenumerals 3 and4 designate the coupled tunable signal circuits feedingthe amplifier 2, and it is to be understood that these tunable circuitshave a band width which is equal at least to twice the maximum frequencydeviation of the F. M. wave. By way of illustration, it is assumed thatthe band width is 200 kc., since F. M. waves may have a permissiblemaximum frequency deviation of i100 kc. The amplified signals are thentransmitted to a first detector network 5 through a second tunable bandpass network which comprises the coupled circuit 6 and l. This latterband pass network, also, has a band width of 200 kc.

The locally produced oscillations which are heterodyned with the signalenergy to provide the I. F. energy, are produced by a local oscillatorwhich comprises the tube 8 having a tunable tank circuit 9. The cathodeof the local oscillator tube is returned to a tap on the tank coil, andthe local oscillator itself is generally conventional in construction.It will be understood that the tunable tank circuit 9 is varied througha frequency range which is generally higher than the signal frequencyrange by the I. F. value at all corresponding points of the two ranges.

For the purposes of this application the I. F. value is assumed to be456 kc., which is the magnitude of the I. F. generally employed forreceiving signals in the standard broadcast range of 55C-1700 kc. Theletter M denotes the coupling between the first detector circuit 5 andthe local oscillator. It is to be understood that the local oscillationsmay be injected into the first detector circuit in any desired and wellknown manner. The numeral l0 designates the usual mechanical uni-controlmechanism which adjusts the tuning elements of the oscillator tankcircuit 9 and the various signal circuits 3, 4, 6 and 1, and thiscontrol device is represented by the dotted lines I0.

'Ihe numeral II designates the resonant output circuit of the rstdetector, and, as stated previously, this circuit is assumed to be tunedto a center I. F. value of 456 kc. This assumed value is, of course, thecase where it is desired to employ the same I. F. networks for receivingthe amplitude modulated waves in the standard broadcast range. In thiscase the various signal networks and the local oscillator tank circuitwould be adjusted so as to receive standard broadcast signals. Where,however, solely F. M. waves are to be received, and it may be desired toreceive F. M. waves having a frequency deviation of kc. or 50 kc., thenit would only be necessary to vary the band width of the coupledcircuits 3-4 and 6 1. In this latter case, the operating I. F. value maybe of the order of 3 mc., and, therefore, the oscillator frequency rangewould be appropriately chosen to provide such an I. F. value. However,for the ypurposes of the present application let it be assumed that thereceiving system, as shown in Fig. 1, is the result of adjustment toimpress F. M. waves having a frequency deviation of |100 kc. on asuperheterodyne receiver whose I. F. networks are of a band width of 50kc. and wherein the I. F. value is 456 kc.

'I'he I. F. amplifier tube I2 is of the usual construction, and has itssignal grid coupled to the high potential side of the resonant inputcircuit I3, the latter being tuned to the operating I. F. value.Circuits II and I3 are reactively coupled so as to provide an effectiveband width of 50 kc. The low potential side of input circuit I3 may beconnected to the grounded end of the grid bias resistor I4. A commonsource of positive potential may be provided for the screen grid and theplate of tube I2. The numeral I5 designates a resistive load arranged inthe plate circuit of the I. F. amplifier.

An aperiodic amplifier follows the tuned I. F. amplifier I2. Theaperiodic amplifier comprises a tube I6 whose signal grid is coupled tothe plate end of resistor I5 through a direct current blocking condenserIl. The signal grid is connected to the grounded end of the grid biasresistor I8 by the grid leak resistor I9. A common source of positivepotential supplies both the screen grid and plate I6, and the numeral 2Udesignates the resistive load in the plate circuit of tube I6. Due tothe fact that in a 50 kc. double-tuned bandpass stage the time delay,assuming an phase shift over the band, would be equal to, or largerthan, l0 microseconds, bandpass couplings are used only in two stages.network is used between the first detector and the first I. F.amplifier, and the second bandpass coupling network is used between thelimiter tube and the second detector tube. Between the other I. F.stages resistance couplings are utilized.

The capacities and resistors for these resistance-coupled I. F. stagesare chosen to produce a very broad characteristic with maximum gain atapproximately 456 kc. Tubes of the GSK? type may be used in theseapericdic stages, and it is to be understoodl that more than oneaperiodic amplifier stage may be employed. A gain of A bandpass couplingapproximately Sli-.per stage can be obtained with a time delay of lessthan one microsecond per stage. The total time delay of the I. F.amplifier network, measured between the first detector and the seconddetector input, may be approximately 22 microseconds.4 In other words,inherent time delay occurring in the I. F. network may be minimized byemploying only as many resonant I. F. circuits as may be required forselectivity purposes, while the remaining I. F. gain is obtained withresistance-coupled circuits, or with other low time delay I. F. stages.

Prior to detection of the F. M. signals, the latter are passed through alimiter stage. The limiter employed in the receiving system utilizes atube of the SSK'I type, and which tube is designated by the numeral 2 l.The signal grid of the tube is coupled to the resistor element by thedirect current blocking condenser 22, and the signal grid is connectedto the ground by the grid leak resistor 24. A common source of positivepotential is connected to the screen grid and plate of the limiter tube.The resonant primary circuit of the second I. F. bandpass network isdisposed in the plate circuit of tube 2|.

The second bandpass network comprises the primary circuit 215 and thesecondary circuit 25. The symbol M1 denotes the reactive couplingbetween these circuits which provides a bandpass network with a bandwidth of kc. Each of the primary and secondary circuits 25 and 2,6 is,of course, resonated to the I. F. Value, and the high potential side ofcircuit 25 is connected to the midpoint of the coil of circuit 26through the condenser 21.

The constants of limiter 2| are so chosen that the tube functionstolimit any amplitude Variations in the F. M. signals. The grid condenser22 and resistor 24 are so chosen as to permit developed grid bias tofollow the variations in F. M. grid signal amplitude. For example, vatime constant of 5 microseconds could be used for 22-24, since suchvalue would permit following any audible audio frequency.

The limited I. F. signals are impressed upon The resultant voltageacross resistorv 3| is ernployed as the audio modulation voltage, and istransmitted to one or more audio amplifier networks.

The discriminator network 2'5--26 has a peak separation of 50 kc. andslope of 0.3 volt per kc.

maximum. The output of the discriminator is detector network shown inFig. 1 has been disclosed and claimed by S. W. Seeley in U. S. P.

2,121,103, granted June 21, 1938. Particular reference is made to Fig. 5of the said Seeley Ypatent which shows a second detector network of thepresent type. From the ungrounded end of the output resistor 3| of thedetector there is derived the audio voltage which corresponds to thefrequency deviations of the F. M. wave. It is not believed necessary torecapitulate in the present description the functioning of the seconddetector network, since those skilled in the art are fully aware of themanner in which such a network functions. It is believed sufficient topoint out that the frequency detector shown has the characteristicpresented in Fig. 2. The discriminator characteristic slope directiondepends on many factors. It is important, however, that it nally be suchthat the phase of audio feedback Voltage causes the frequency deviationenvelope of the oscillator to be in phase with that of the F. M.signals. From this characteristic it is seen that as the frequency ofthe F. M. wave swings to and fro with respect to the center frequency(fc) of the I. F. band, there flows through the output resistor auni-directional current whose-magnitude and polarity depends upon themagnitude and direction of frequency deviation.

' denser 3l at l5 kc.

fed through a. two-section, low-pass filter 32 which has a cut-offfrequency of 200 fkc. and a time delay of approximately 3 microseconds.The audio output of the filter 32 passes to the audio utilizationnetwork. It is also fed through the degenerative audio feedback 'path33. The reason for the selection of a cut-olf frequency for lter 32which is as high as 200 kc., is the desirability of having a small timedelay. The lower the cut-oif frequency of a filter, the higher is thetime delay. Hence, a Value is chosen which minimizes time delay and isconsistent with adequate filtering of the I. F.

The audio voltage fed through the feedback path 33 is impressed upon atube 34 which is included in a phase correction stage. This phasecorrection stage provides a phase shift approximately equal in magnitudebut opposite in phase to that existing between vthe input signal and thesignal in the feedback path. This phase advance is accomplished byarranging an inductance 35 and resistance v315 in series in the platecircuit of tube 34. The ratio of the inductance of coil 35 to theresistance of resistor 36 is chosen to be equal to 28 microseconds. Inorder to cut down the gain of this correction stage for high frequency,the inductance 35 is resonated by coninput signal grid coupled to theline33 through condenser 3'8, while the grid is also connected to thegrounded end of the bias resistor 39 through the grid leak resistor 40.The plate of the tube S4 is connected to a source of positive potentialby a resistor 4| of approximately 10,000 ohms, and the direct currentblocking condenser-G2r connects the upper end of coil 35 to the plate oftube 34. On a steady state basis the stage 34 should compensate forphase delay caused in the I. F. networks, the discriminator and in theaudio filter network.

The output signal energy of the correction stage is applied to the inputgrid of an electronic reactance tube 50. The function of this tube is toprovide across the tank circuit 9 of the local oscillator a reactiveeffect which will vary the frequency of the oscillator circuit. Thereactance tube circuit appears as a negative inductance across the tankcircuit 9. This is accomplished by connecting the plate of theoscillator tube 8 to ground through a path which includes the seriescondensers 5I and 52.` 'Ihe cathode of tube 50 is connected to groundthrough a bias resistor 53, while the grid 54 is connected to groundthrough the grid leak resistor 55. The grid-54 is connected to thejunction of condensers 5I-52, and hence alternating voltage dvelopedacross condenser 52 is impressed on grid 54.- 'I'he input grid of tube50 is denoted by the numeral 55'., and the latter may be surroundedby apositive screening eld. The plate 60 of tube 50' is connected to asource of positive potential through a radio frequency choke coil, andthe plate is also coupled to the high potential side of tank circuit 9through the condenser 70. The tube 50 may be of 6L? type, while theoscillator tube 0 may be of 1852 type.

It can be'readily demonstrated that tube 50 appears asanegativeinductance acrosstankcir- The correction stage has its oscillatorfrequency at an audio rate.

inductance is inversely proportional to the gm (mutual conductance) oftube 50.

When the audio voltage on grid 55 of tube 5U swings toward the positivedirection the reflected negative inductance across tank circuit 9decreases. Since the reflected inductance is in shunt with the positiveinductance of tank circuit 8, it reduces the total inductance in thecircuit and the oscillator frequency decreases.

Thus, for example, if the audio frequency voltage applied to grid 55' isof 400 cycles, the effective inductancemegative) produced by the tube 50changes amplitude in accordance with instantaneous amplitude changes ingrid 55' four hundred time a second. The resultant frequency change inthe oscillator may be readily computed from a knowledge of the effectiveL and C of the tank circuit. Thus, if the mutual conductance of tube 50increases simultaneously on a positive audio modulation peak, theeffective L of the tank circuit increases and the frequency decreases.

Considering the action more specifically, across condenser 52 there isdeveloped a control voltage which is in quadrature with the voltageacross the tank circuit 9. This quadrature voltage is impressed on grid54, and tube 50 feeds back into the tank circuit 9 through condenser 10alternating current which is in quadrature with the tank circuitvoltage. The grid 55' is varied at the rate of the audio voltageimpressed on it through the audio feedback path, and, therefore, thesimulated reactive effect across tank circuit 9 is varied at the ratedetermined by the fed back audio voltage. That is, the quadraturecurrent fiowing into the tank 9 is varied in amplitude at an audio rate,and changes the As a result of this arrangement the local oscillationswhich are fed to the first detector are frequency modulated. 'Ihefrequency deviation provided is 150 kc. total, or l75 kc. on either sideof the unmodulated oscillator frequency. The frequency modulated localoscillations which are fed into the rst detector are arranged so thatits frequency deviation envelope is in phase with that of the F. M.signals impressed on the first detector at circuit 1. Hence, when thetwo signals beat in the first detector the resultant total I. F. peak topeak frequency deviation is reduced to the difference of 200 kc. and 100kc.

As a result there is developed across the I. F. output circuit Il F. M.waves which are at the I. VF. value, but whose frequency deviation has amaximum of $25 kc. In other words the maximum frequency deviation of theresulting I. F. signals is half the band width of the I. F. networks.The frequency modulated oscillator voltage is injected into the firstdetector in any desired manner.

It will now be observed that the phase correction stage 36 counteractsthe infiuence of the time, or phase, delay in the I. F. circuits and thephase lag between the envelopes of the frequency deviation inthe inputsignal and in the heterodyned oscillator output voltage. The correctionstage functions to advance the phase in the feedback path between theaudio detector andthe oscillator reactance control tube. In designingthe phase advancing stage, it is necessary to cut off any feedback atfrequencies which produce a net phase shift of 90 degrees or more. Thisprecaution prevents regenerative feedback at such high frequencies. Itmay be desirable to employ cuit 9, and that the magnitude of suchnegativeI more than one phase correcting stage prior to the reactancetube 50 in order to compensate for the time delay which is added byutilization of the low pass filter 32. Generally speaking, however, thephase correction stage in the audio feedback path should substantiallycorrect for the time delay occurring up to the audio detector, whilehaving a sharp cut-off characteristic so as to prevent oscillation atthe high audio frequencies.

While we have indicated and described a system for carrying ourinvention into effect, it will be apparent to one skilled in the artthat our invention is by no means limited to the particular organizationshown and described, but that many modifications may be made withoutdeparting from the scope of our invention, as set forth in the appendedclaims.

What we claim is:

l. In a method of receiving frequency modulated carrier waves,heterodyning collected frequency modulated waves with local oscillationsof a frequency different from the frequency of the collected Waves by anoperating intermediate frequency, deriving from the intermediatefrequency waves modulation component frequency voltage which correspondsto the frequency modulation of the collected Waves, and varying thefrequency of the local oscillations with the said modulation voltagethereby to produce frequency modulated local oscillations having afrequency deviation which is differentfrom the frequency deviation ofthe aforesaid collected Waves, amplifying said intermediate frequencywaves whereby a phase delay occurs, and advancing the phase of theaforesaid audio voltage prior to frequency modulation of said localoscillations to an extent sufficient to compensate for said phase delay.

2. In a method of receiving frequency modulated carrier waves,heterodyning collected frequency modulated waves with local oscillationsof a frequency different from the frequency of the collected waves by anoperating intermediate frequency, deriving from the intermediatefrequency waves modulation component frequency voltage which correspondsto the frequency modulation of the collected Waves, and varying thefrequency of the local oscillations with the said modulation voltagethereby to produce frequency modulated local oscillations having afrequency deviation which is different from the frequenny deviation ofthe aforesaid collected waves, and advancing the phase of the aforesaidaudio voltage prior to frequency modulation of said local oscillationsto an extent sufficient to maintain the envelope of said collected Wavesand the envelope of said frequency modulated local oscillations inphase.

3. In a receiving system of the type adapted to receive frequencymodulated rcarrier waves, a first detector circuit having an inputcircuit upon which such waves are impressed, said input circuit having aband width which is relatively wide, a demodulator network, a networkcoupling said first detector to said demodulator, said coupling networkbeing tuned to an operating intermediate frequency and having arelatively narrower band Width than said first mentioned band Width, alocal oscillator network' arranged to produce local oscillations whichdiffer from said impressed frequency modulated waves by saidintermediate frequency value, means for controlling the frequency ofsaid local oscillator network, said last means comprising an electronicdevice connected to said oscillator network to provide an effectivereactance control therefor, and a degenerative modulation voltagefeedback path between the demodulator output and said electronic devicewhereby the local oscillations are varied in frequency over a frequencyrange which diiers from the frequency range of said relatively wide bandWidth by the value of said relatively narrower band Width, and means forimpressing said modulated local oscillations upon said first detector

